The invention relates to a method for controlling two electrically series-connected reverse conductive IGBTs of a half bridge, at which a DC operating voltage is present, wherein these reverse conductive IGBTs have three switching states “+15V”, “0V” and “−15V”.
Reverse conductive IGBTs are also known as RC-IGBTs. An RC-IGBT differs from a conventional IGBT in that the diode function and the IGBT function are combined into one chip. The result is a power semiconductor in which the anode efficiency in the diode mode is dependent on the presence of a gate voltage.
The basic structure of an RC-IGBT is shown in greater detail in cross section in FIG. 1. This structure is known from publication “A High Current 3300V Module Employing Reverse Conductive IGBTs Setting a New Benchmark in Output Power Capability” by M. Rahimo, U. Schlapbach, A. Kopta, J. Vobecky, D. Schneider, A. Baschnagel, published in ISPSD 2008. This basic structure consists of a lightly n-doped substrate Sn, which is provided on the collector side with an n-doped layer Fs. Applied to this layer Fs is a highly-doped p-layer Sp which for its part is provided with a metal layer MK. Disposed in this highly-doped p-layer Sp are highly-doped n-areas Ln, such that said areas lie in the shadow of highly-doped p areas Wp let into the lightly n-doped substrate Sn. These are embodied such that these each form a so-called trough around a recessed area of a metal layer ME, which serves as emitter gate of the RC-IGBT. These recesses each break through a metal layer MG, which in relation to the metal layer ME in each case, which serves as emitter gate of the RC-IGBT, and compared to the lightly n-doped substrate Sn, is surrounded by a silicon oxide layer. In addition each recess of the metal layer ME serving as an emitter gate is surrounded in the trough-shaped, highly-doped p-area Wp by a highly-doped n-layer Sn+.
With a gate emitter voltage below a threshold voltage of the MOS channel (15V) of a reverse conductive IGBT the anode efficiency is high, whereby the charge carrier density in the conductive state is high and the conductive state voltage is low. By contrast the reverse-recovery charging, the reverse-recovery losses and the switch-on losses of an RC-IGBT opposite the bridge branch are high. With a gate emitter voltage above a threshold voltage (+15V) of the MOS channel of a reverse conductive IGBT the anode efficiency is low, whereby the charge carrier density in the conductive state is low and the conductive state voltage is high. Since the MOS channel is switched on, this RC-IGBT cannot accept any blocking voltage.
Because of this fact an activation, and thus a method for controlling a conventional IGBT, cannot be used with a reverse conductive IGBT. How a method for controlling an RC-IGBT can look is to be found in the publication already mentioned. It is characteristic of this method that the switching state of the reverse conductive IGBT depends not only on a required value of an output voltage of a multi-phase current converter with RC-IGBTs as current converter valves, but also on a direction of current of the collector current.
FIG. 2 shows an equivalent circuit diagram of a bridge branch 2 of a current converter, wherein the RC-IGBTs T1 and T2 are used as current converter valves. This bridge branch 2, also referred to as a half bridge, is connected electrically in parallel by means of two bus bars 6 and 8 to a direct current source 4. The two reverse conductive IGBTs T1 and T2 of the bridge branch 2 are connected electrically in series. A connection point of these two reverse conductive IGBTs T1 and T2 forms an AC voltage-side terminal A, to which a load is able to be connected. The DC voltage source 4 has two capacitors 10 and 12, which are likewise connected electrically in series. A connecting point of these two capacitors 10 and 12 forms a midpoint terminal M. A DC voltage Ud is present at these two capacitors 10 and 12 connected electrically in series. As an alternative, instead of the two capacitors 10 and 12, just one capacitor can also be used, which is disposed between the two bus bars 6 and 8. The midpoint M is then no longer accessible. With an intermediate voltage converter this DC voltage source 4 forms an intermediate voltage circuit, wherein the DC voltage Ud present is then referred to as the intermediate circuit voltage. The bridge branch 2 is present three times in a three-phase converter, especially a pulse converter, which is used as a load-side converter of an intermediate voltage circuit converter. A pulse-width modulated square-wave voltage UAM is present relative to the midpoint terminal M of the DC voltage source 4 at the AC voltage-side output A.
A block diagram of a control and regulation device of a three-phase current converter, especially a pulse current converter of an intermediate voltage circuit converter with the associated semiconductor-like activation facilities 14 of a bridge branch 2 of this current converter, is shown in FIG. 3. A control device 16 generates two desired control signals S*T1, S*T2, S*T3, S*T4, S*T5 and S*T6 as a function of a desired value, for example a rotational speed desired value n*, for each bridge branch. For reasons of clarity only the bridge branch 2 of the three bridge branches of a three-phase current converter is shown. The two desired control signals S*T1 and S*T2 are each fed to a semiconductor-like activation facility 14 of each reverse conductive IGBT T1 and T2 of the bridge branch 2. On the output side an actual control signal ST1 or ST2 is present in each case, with which a gate G of a respective reverse conductive IGBT T1 or T2 is activated. In this diagram the AC voltage-side terminal of the bridge branch 2 is not labeled with the letter A, as in the diagram depicted in FIG. 1, but with the letter R. The three bridge branches of a three-phase current converter are connected to each other by means of the two bus bars 6 and 8 and are connected electrically in parallel to the DC voltage source 4.
As already mentioned the stationary switching state of the two reverse conductive IGBTs T1 and T2 of a bridge branch 2 is not only dependent on the desired value of an output voltage u*AM, but also on the polarity of an output current iA of this bridge branch 2. Whenever the reverse conductive IGBT T1 or T2 is to conduct current in the reverse direction (negative collector current, diode mode) it is switched off. In this way the charge carrier concentration in diode mode is raised. The switching states of the two reverse conductive IGBTs T1 and T2 of the bridge branch 2 can be taken from the following table:
UAO(targ.)iA(act.)T1T2+Ud/2>0onoff+Ud/2<0offoff−Ud/2>0offoff−Ud/2<0offon
FIGS. 4 to 8 respectively show signal waveforms plotted over time t in a diagram for the case in which, for the negative polarity of the output current iA, the reverse conductive IGBT T1 is operated in diode mode and the reverse conductive IGBT T2 is operated in IGBT mode. FIG. 4 shows the waveform of the desired output voltage u*AM over time t. To enable this desired output voltage u*AM to be converted, the desired control signals S*T1, and S*T2 are needed, the temporal waveforms of which are shown, plotted over time t, in FIGS. 5 and 6.
At point in time t0 the value of the desired output voltage u*AM is equal to half the value of the DC voltage Ud present at the DC voltage source 4. This makes the reverse-conductive IGBT T1 operated in diode mode current-conductive. So that this reverse conductive IGBT T1 can conduct current in diode mode, this IGBT must be switched off. In the signal waveform of the diagram according to FIG. 7, which shows the waveform of the gate voltage uGE(T1) of the reverse conductive IGBT T1 over the time t, the gate voltage is in the off state (−15V). At this point in time the switching state of the reverse conductive IGBT T2 in accordance with the gate voltage uGE(T2) corresponding to FIG. 8 is likewise in the off state. At point in time t1 the desired output voltage u*AM changes from +Ud/2 to −Ud/2. At this point in time t1 the desired control signal S*T1 changes from high to low, whereas the desired control signal S*T2 changes from low to high. Through this the reverse conductive IGBT T1 operated in diode mode is switched on for a predetermined period of time ΔT1, wherein the IGBT T2 operated in IGBT mode remains in the switched-off state. In this predetermined period of time ΔT1 the reverse conductive IGBT T1 operated in diode mode continues to be current-conductive, whereby the charge carrier concentration falls.
At point in time t2 the reverse conductive IGBT T1 operated in diode mode is switched off again. After a further predetermined period of time ΔTV has elapsed, which is also referred to as the blocking time, at point in time t3 the reverse conductive IGBT T2 operated in IGBT mode is switched on (FIG. 8). The commutation from the IGBT T1 operated in diode mode to the reverse conductive IGBT T2 operated in IGBT mode occurs at point in time t3. The blocking time ΔTV is necessary in order to avoid a bridge branch short-circuit. This blocking time ΔTV again leads however to a rise in the charge carrier concentration in the reverse conductive IGBT T1 operated in diode mode and thus to an increase in the switching losses.
At point in time t5 the desired output voltage u*AM changes again from −Ud/2 to +Ud/2. The reverse conductive IGBT T2 operated in IGBT mode remains switched on for a predetermined period of time ΔT2 calculated from point in time t5, before this IGBT is switched off at point in time t6 and the output current iA commutes to the reverse conductive IGBT T1 operated in diode mode.
So that the effect of the anode efficiency works, the blocking time ΔTV should be as small as possible between the switching off of the reverse conductive IGBT T1 operating in diode mode and the switching on of the reverse conductive IGBT T2 operating in IGBT mode. The control method (commutation method) disclosed in the publication cited at the start is time-controlled, which requires high timing precision. If the anode efficiency is high, the conductive voltage is low so that the reverse recovery losses are reduced.
The underlying object of the invention is now to further develop the known method for controlling a reverse conductive IGBT such that the reverse-recovery charge becomes as low as possible in combination with a conductive voltage which is as low as possible, and that in diode mode a high surge withstand strength is achieved.